Microphone with voltage pump

ABSTRACT

An integrated circuit ( 102 ) configured to provide a microphone output signal, comprising: a preamplifier ( 108 ) coupled to receive an input signal, generated by a first microphone member that is movable relative to a second microphone member; and a voltage pump ( 104 ) to provide a bias voltage to either microphone member.

This invention relates to an integrated circuit with a voltage pump anda microphone preamplifier.

BACKGROUND

For many years the preferred type of microphone for telecom applications(ie Mobile phones) has been electret microphones. This type ofmicrophone is based on the principle of a capacitor which is formed by amovable member that constitutes a membrane of the microphone and anothermember, eg a so-called back plate of the microphone. One of the membersof the microphone, preferably the membrane, is provided with a capturedelectrical charge also known as an electret layer.

However, in recent years microphones are also provided with the chargeby a DC voltage source during operation. These microphones compriseordinary condenser microphones and MEMS microphones which are on theirway to be utilized in telecom applications.

Microphones without an electret layer can be implemented in twodifferent ways. That is, on the one hand, in the conventional way wherethe parts are manufactured as metal parts for mounting in a case (whichnormally forms both a capacitor plate of the microphone and abackplate).

On the other hand, in recent years also silicon usually used forintegrated circuits have been used for the manufacture of mechanicalstructures. This technology is usually denoted MEMS. The differencesbetween manufacture of microphones in MEMS technology and conventionaltechnology are mainly the processes involved and the tolerances. I.e.MEMS technology requires clean rooms and silicon technologies. Theprecision is higher for silicon technologies but so is the cost.

The characteristics of the two types of microphones are that themicrophone capacitances are usually smaller and the required biasvoltages are smaller. This is due to the fact that microphonesimplemented in silicon have to be smaller to be able to competecost-wise. This means that the silicon microphones are optimizeddifferently even though the principle of the two microphone types is thesame. I.e. the membrane area of a silicon microphone is typically 1mmˆ2, airgap 1-2 um, capacitance 1 pF and bias voltage 10V. And for aconventional microphone it is area=3 mmˆ3, airgap=10-15 um, capacitance3 pF and bias voltage 30-40V. Or even larger bias voltage.

As the charge on the microphone capacitor has to be kept constant tomaintain proportionality between sound pressure and voltage across thecapacitor members, it is important not to introduce any de-charging ofthe microphone.

Therefore, in order to pick up a microphone signal from the capacitor,amplifiers configured with the primary objective of providing high inputresistance are preferred to buffer the capacitor from circuits which areoptimized for other objectives. The amplifier connected to pick up themicrophone signal is typically denoted a preamplifier or a bufferamplifier or simply a buffer. The preamplifier is typically connectedphysically very close to the capacitor—within a distance of very fewmillimetres or fractions of millimetres.

For small-sized microphones only a very limited amount of electricalcharge can be stored on one of the microphone members. This furthers therequirement for high input resistance. Consequently, the inputresistance of preamplifiers for small-sized microphones has to beextremely high—in the magnitude of Giga ohms. Additionally, the inputcapacitance of this amplifier has to be very small in order to achieve afair sensitivity to sound pressure.

Telecom microphones with an integrated preamplifier are sold in highvolumes and at very low prices. As the cost of an amplifier for atelecom microphone is directly related to the size of the preamplifierchip die, it is important, for the purpose of reducing price, that thepreamplifier die is as small as possible.

So, obviously, there is a need for microphone preamplifiers with gainand very low input capacitance, and the lowest possible preamplifier diearea. Additionally, low noise is important. Low noise is important asnoise can be traded for area—ie if the circuit has low noise and a noiselower than required, this noise level overhead can be traded for lowerchip die area and it is thus possible to manufacture the preamplifier atlower cost.

However, since sensitivity is traded for low prise, microphones fortelecommunications purpose are less sensitive. From a marketperspective, there is a demand for a larger sensitivity of themicrophone and preamplifier in combination. So therefore the gain in thepreamplifier is to be increased to meet the demand. Additionally, thereis a demand for low noise in the audible range. Moreover, in order toensure a good signal-to-noise ratio while meeting the demand for arelatively large sensitivity, the input capacitance of the preamplifiermust be small to avoid an unnecessary signal loss from the microphone(cf. the equivalency of the microphone signal being exposed to a voltagedivider constituted by the capacitances).

Since the chip area occupied by the preamplifier must be as small aspossible to obtain relatively low cost, the preamplifier must be assmall as can be. Therefore, since amplifier configurations known fromhearing aids are generally not optimised for chip area to the sameextent, these configurations are not applicable. Further, one shouldbear in mind that buffers or amplifiers applied in hearing aids are notconfigured to provide such high gain levels as are required for thelow-sensitivity microphones used in telecommunication applications. Inhearing aids chips more space is required for the same noise performancesince buffers are required to avoid overload in hearing aids.

The charge on the microphone capacitor can be provided by a relativelyhigh DC supply voltage or by a manufacturing process where a staticcharge is captured on one of the capacitor members eg on the membranewhich can made from Teflon. The type of microphone where a static chargehas been applied is preferred in telecom applications since this typedoes not require a circuit for supplying the charge by a DC voltage.However, this type has been found shown to lose the charge when exposedto relatively high temperatures. Additionally, such microphones requirecareful handling and mounting during eg a soldering process whichinherently exposes the microphone to high temperatures. When themicrophone has lost its static charge, the microphone's ability as asound transducer is diminished and it will be far too inexpedient tore-establish the charge.

Noise is also an important parameter for telecom microphones. Typically,the predominant noise sources are related to the preamplifier in themicrophone. But when switching and/or oscillating circuits areincorporated on the chip die, surrounding/neighbouring circuits/circuitpaths with transient signals can become major noise sources.Additionally, when a noisy signal is transmitted on the same path orterminal as another signal, eg the microphone signal, this noisy signalwill constitute a major and direct noise source of which the noiseinfluence may be difficult to suppress.

When designing a preamplifier in CMOS technology for a microphone thereis normally three noise sources. These sources are noise from a biasresistor, 1/f noise from an input transistor, and white noise from theinput transistor. We assume that input transistor noise dominates. Bothwhite noise and 1/f noise can be minimized by optimization of the lengthand the width of the input transistor(s). This applies for any inputstage, eg a single transistor stage or a differential stage. The noisefrom the bias resistor can also be minimized. If the bias resistor ismade very large the noise from the resistor will be high pass filteredand the in-band noise will be very low. This has the effect, though,that the lower bandwidth limit of the amplifier will be very low. Thiscan be a problem as the input of the amplifier will settle at a nominalvalue only after a very long period of time after power-up.Additionally, signals with intensive low frequency content arising formeg slamming of a door or infra sound in a car can overload theamplifier. Another related problem is small leakage currents originatingfrom mounting of the die inside a microphone module. Such currents willdue to the extreme input impedance establish a DC offset. This willreduce the overload margin of the amplifier.

PRIOR ART

US 2003/0235315-A1 discloses a digital microphone comprising an electretcondenser microphone, a preamplifier, a limiter and a sigma-deltamodulator to provide a digital output bit stream which is responsive toa sound pressure on the electret microphone. The sigma-delta modulatorprovides a single bit stream output at a high (oversampled) bit rate.The preamplifier, limiter, and sigma-delta converter is implemented onan integrated circuit using larger geometry analog IC technology. Themicrophone is utilized in a cellular phone.

US 2002/0071578 discloses a microphone comprising in its housing anelectret condenser microphone and a preamplifier of the JFET type. Thepreamplifier is integrated on a first integrated circuit and provides amicrophone output signal to a second but external integrated circuit.The output signal is coupled to an input integrator of a sigma-delta A/Dconverter on the second but external integrated circuit. The inputintegrator provides biasing for the JFET.

SUMMARY OF THE INVENTION

There is provided an integrated circuit configured to provide amicrophone output signal, comprising: a preamplifier coupled to receivean input signal, generated by a first microphone member that is movablerelative to a second microphone member; and a voltage pump to provide abias voltage to either microphone member.

Consequently, since sensitivity a microphone is closely related to thecharge provided on one of the microphone members a microphone outputsignal which is more sensitive to a sound pressure on one of themicrophone members is provided. In combination therewith the microphoneoutput signal is provided by a preamplifier which provides the outputsignal as a buffered output signal with a desired gain.

Since the preamplifier and the voltage pump is embodied as a integratedcircuit on a semiconductor substrate, close integration of a microphoneelement and the integrated circuit is possible. Additionally, theversatility of a microphone integrated with the preamplifier and thevoltage pump is increased since other circuit systems, providingelectrical operating conditions for the microphone and further signalprocessing of the microphone output signal, does not need to provide ahigh bias voltage to the microphone. This is a major improvement sincebias voltages are significantly higher than common nominal voltagelevels of integrated circuit technologies. Bias voltages can be as highas 10, 20 up to around 60 volts, whereas common nominal voltage levelsare about 3 or 5 volts.

Expediently, the integrated circuit is configured with an oscillatordriven voltage pump to provide a bias voltage to either microphonemember; and the oscillator is configured to draw substantially equallevels of current across signal cycles provided by the oscillator.Thereby, especially when the microphone output signal and the operatingpower is provided via a shared terminal (and path), the quality of themicrophone signal is improved. This is achieved since the oscillatordraws a more constant level of current. Otherwise the microphone outputsignal would be level-shifted as the oscillator would draw differentlevels of current from half-cycle to half-cycle. Additionally, theoscillator will generate less switching noise.

In an expedient embodiment, the oscillator comprises paths with elementsthat can be charged with an electrical charge and where the paths arecontrolled by the oscillator to charge the different elements of thedifferent paths alternately by a current drawn from a common source. Theelements, in the form of capacitors, are charged alternately and coupledto level sensitive triggers for providing 180 degrees phase shifted,square formed oscillator signals.

Preferably, the voltage pump has a first pump stage at which anoscillating signal with a voltage pulse level is pumped to a highervoltage pulse level, and a second pump stage at which a voltage level ispumped to a higher level by means of a circuit operating on theoscillator signal, provided at the first stage, with the higher voltagepulse level. Thereby an area efficient voltage pump is provided.Especially for telecom microphones this is an important parameter.

The integrated circuit can comprise a first portion configured with acircuit component layout for electrical operation at or below a nominalvoltage level and a second portion configured with a circuit componentlayout for electrical operation above the nominal voltage. The firstportion is also denoted a low voltage section and the second portion isdenoted a high voltage section. Components in the high voltage portionare larger than the components in the low voltage portion. Thecross-portion implementation of the voltage pump provides for more chiparea efficient implementation of the pump.

When the pulse amplitude of oscillator signals provided at the lowvoltage section are substantially equal to the nominal voltage of thelow voltage section, the high voltage part can be configured with simplevoltage pumps. This gives the advantage of enabling implementation ofthe pump using cost efficient parasitic components which can withstandthe high voltage.

Generally, in an integrated circuit it is cumbersome to provide feedbackinvolving voltages substantially above a nominal operating voltage.Therefore, in an expedient embodiment, the output signal of the firstvoltage pump stage is provided as a feedback signal to a circuit whichprovides a regulated voltage pulse level of the signals output from thefirst pump stage. Thereby, it is possible to provide a relativelyprecise bias voltage by means of a feedback configuration operating alower voltages.

Preferably, the voltage pump has a first pump stage providing anintermediate bias voltage and a second pump stage providing the biasvoltage from the intermediate bias voltage; and the second pump stagecomprises a voltage pump configured as a Dickson converter. The Dicksonconverter provides low noise, low ripple on its output signal when ithas reached a stable state and is capacitively loaded (by a microphone).Further, it has a simple circuit configuration and requires small areafor implementation.

Expediently, an output signal of the voltage converter of the Dicksontype is provided as a feedback signal to a circuit which provides aregulated voltage pulse level of oscillator signals operating thevoltage converter. Thereby, a simple feedback configuration is provided.

When multiple voltage converters are cascaded very high voltages (abovee.g. 20 or 30 volts) can be provided. However, since the multiplevoltage converters are simple in their configuration in order to providechip area and cost efficient voltage pumps, the bias voltage can varysignificantly from chip die to chip die. Thus, preferably multiplevoltage converters are cascaded to provide the bias voltage, and afurther voltage converter, which matches the first converter in thecascade, is coupled to receive the same signal as the first converterand to provide a feedback signal to a circuit which maintains a fixedvoltage level of the signals output from the further voltage converter.Thereby the further converter and the first converter is coupled as amaster slave configuration. Since the master/slave converters can bemade almost identical a feedback loop enclosing a portion of the cascadeis provided. Thereby, provided bias voltages can be controlled muchbetter from chip die to chip die.

The integrated circuit can be implemented by various IC technologies,however, high voltage components required for voltage pumps requirelarger spacing, deeper-wells, thicker gate oxide etc. Technologies withsuch components exists but they are expensive and are generally notavailable for low cost telecom applications. But when the voltage pumpcomprises capacitors implemented as Metal capacitors, standardtechnologies can be utilized. These components are based on parasiticsand are thus not very precise and well controlled. But the use of suchcomponents is successful if the simple Dickson multiplier isimplemented.

Correspondingly, in a preferred embodiment the voltage pump comprisesdiodes implemented as Poly-diodes.

Still correspondingly, in a preferred embodiment the voltage pumpcomprises diodes implemented as diffusion diodes in an N-well.

The intrinsic sensitivity of telecom microphones are normally relativelylow despite being provided with a relatively high bias voltage. As aconsequence of this the preamplifier requires gain. Furthermore highsensitivity is required. The consequence of this is that higher gainshould be provided by the preamplifier. However, it is desired toprovide high overload margins and the ability to handle large lowfrequency signals such as car rumbling and door slamming. Additionally,low frequency signals can comprise pulses generated by start-up of thevoltage pump.

To meet these demands a preamplifier is provided which comprises adifferential input stage with a first and a second input terminal and anoutput stage with an output terminal; a feedback circuit, with alow-pass frequency transfer function, coupled between the outputterminal and the first input terminal and integrated on thesemiconductor substrate; and where the second input terminal provides aninput for a microphone signal.

Thereby a semiconductor microphone preamplifier is provided with afilter feedback configuration. This preamplifier can provide a largeloop-gain outside the audio band and will give rise to very littledistortion in the audio band. But more importantly, inter-modulationdistortion introduced by frequency components at low frequencies,outside the audio band, will be very low. The loop-gain characteristicprovided by the feed-back configuration provides e.g. lower distortion.

Preferably, the feedback circuit is a filter with a transfer function,in the frequency domain, with a zero and a pole; wherein the zero islocated at a higher frequency than the pole.

Expediently, the preamplifier has a transfer function, in the frequencydomain, with a zero and a pole; wherein the pole is located in the range0.1 Hz to 50 Hz or 0.1 Hz to 100 Hz or 0.1 to 200 Hz.

The feedback circuit can be configured as a filter which, in thefrequency domain, has a relatively high gain level below a transitionfrequency range and a relatively low gain level above the transitionfrequency range.

In an expedient embodiment, the integrated circuit comprises a DCblocking capacitor coupled to diminish a DC voltage at the input of thepreamplifier, which DC voltage originates from biasing the first orsecond microphone member.

Preferably, the integrated circuit further is configured with ananalogue-to-digital converter; and the voltage pump and theanalogue-to-digital converter are driven by a common clock-signal.Thereby an external clock signal provided for reading out digital bitsof the A/D converter can be utilized as a clock signal for driving thevoltage pump.

BRIEF DESCRIPTION OF THE DRAWING

The invention will be described in more detail in connection with thedrawing in which:

FIG. 1 shows a microphone comprising an integrated circuit with avoltage pump;

FIG. 2 shows a microphone comprising an integrated circuit with avoltage pump operated by a constant current drawing oscillator;

FIG. 3 shows a first oscillator with constant current consumption;

FIG. 4 shows a second oscillator with constant current consumption;

FIG. 5 shows a first embodiment of a first stage voltage pump;

FIG. 6 shows a second embodiment of a first stage voltage pump;

FIG. 7 shows a third embodiment of a first stage voltage pump;

FIG. 8 shows a compound voltage pump;

FIG. 9 shows a second stage voltage pump;

FIG. 10 shows a second stage voltage pump with parasitic capacitors;

FIG. 11 shows an IC implementation of a metal oxide capacitor;

FIG. 12 shows an IC implementation of a poly-diode;

FIG. 13 shows an IC implementation of a diffusion diode in an N-well;

FIG. 14 shows a charge pump with N-MOS switches;

FIG. 15 shows a compound charge pump with a master-slave configuration;

FIG. 16 shows a microphone with an integrated circuit with a voltagepump and a preamplifier with a feedback filter;

FIG. 17 a shows a first feedback filter;

FIG. 17 b shows a second feedback filter;

FIG. 18 shows a bootstrap configuration in connection with a detailedview of the preamplifier;

FIG. 19 shows a first embodiment of a microphone comprising anintegrated circuit with a sigma-delta A/D converter;

FIG. 20 shows a second embodiment of a microphone comprising anintegrated circuit with a sigma-delta A/D converter;

FIG. 21 shows a microphone cartridge;

FIG. 22 shows a schematic view of a microphone with an integratedcircuit and a MEMS microphone member.

DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 shows a microphone comprising an integrated circuit with avoltage pump. The microphone 101 is a compound unit with a microphoneelement 109 and an integrated circuit 102. The integrated circuit 102 isimplemented as a single chip die manufactured by a solid-state diffusionprocess on a semiconductor substrate. The microphone element is formedby a movable member, which constitutes a membrane of the microphone, andanother member, thereby allowing the membrane to move relative to theother member in response to a sound pressure acting on the membrane. Asound pressure detected by the microphone will cause the membrane tomove and consequently change the capacitance of the capacitor formed bythe membrane member and the other member. If the charge on the capacitorformed by these two members is kept constant, the voltage across the twocapacitor members will vary with the sound pressure.

The integrated circuit 102, implemented as a single chip die, is closelyintegrated with the microphone element 109. The close integration of themicrophone element 109 and the integrated circuit 102 is provided byintegration of the microphone element and the chip die in a small-sizedmicrophone cartridge.

The microphone 101 comprises two terminals T1 and T2 which is a groundterminal and a combined microphone signal and power supply terminal,respectively. The terminals T1 and T1 are used for coupling themicrophone (situated in a cartridge) with power supply means and furthersignal processing circuits. Such signal processing circuits aretypically integrated as part of a so-called system-on-chip (SOC) deviceusing IC technology.

The microphone comprises a microphone element 109 illustrated by thecapacitor, Cmic, and an integrated circuit 102. The integrated circuit102 comprises three terminals Tc1, Tc2, Tc3 which is a combinedmicrophone signal and power supply terminal, a combined DC voltage levelfor providing a constant charge on the microphone and a microphoneelement signal, and a ground terminal, respectively.

The integrated circuit 102 comprises a voltage up-converter or voltagepump, UPC, 104 e.g. in the form of a so-called Dickson-converter. Thevoltage pump is operated by an oscillator 103 which preferably providesa square-wave oscillator signal to the voltage pump. Other signals, egsine waves or filtered square waves, with lower contents of harmonicsmay be used to obtain lower noise. The oscillator 103 and the voltagepump 104 are powered by a current drawn through terminal Tc1 (T1) whichis coupled to a nominal supply voltage via a series resistor 106, R.

In response to the oscillator signal the voltage pump 204 provides apumped voltage which exceeds the nominal supply voltage. A nominalvoltage is e.g. 3 or 5 volts and a pumped voltage is e.g. 10, 12, 15, or20 volts. However, higher voltage levels (e.g. up to about 60 volts ormore) can be provided as described later. The pumped voltage isconsidered a DC voltage, but it is superposed by (strong) noisecomponents originating from the oscillator signal on which the voltagepump operates.

The pumped voltage is supplied to a first member of the microphoneelement 109 via a series resistor 105, R. The second member of themicrophone element 109 is coupled to the ground reference. Thereby thevoltage pump provides a constant electrical charge on the microphoneelement. Typically, the first member is the membrane of the microphone.

When a sound pressure acts on the membrane, a microphone element signalis generated at the membrane relative to the second member. Themicrophone element signal is superposed on the pumped voltage providedvia the resistor 105, R. In order to reduce the DC voltage load of theinput stage of the preamplifier 108, A1 the capacitor 107, C1 is coupledas a DC block. To avoid an unacceptable signal loss the capacitor C1should have a capacity larger than approximately ten times the capacityof the microphone element 109.

The preamplifier 108, A1 is powered by drawing an operating currentthrough its output via terminal Tc1 (which is coupled to a nominalsupply voltage via a series resistor 106, R). The preamplifier receivesan input signal and provides an output signal at its output. Thepreamplifier is configured as a simple buffer amplifier or as anamplifier providing a gain of more than 0 db. The preamplifier isalternatively configured to provide an expedient frequency dependentgain response; this will be described in the following.

Reverting to the issue of noise generated by the voltage pump 104 andthe oscillator 103 the following applies: As the voltage pump is anelectronic circuit it will generate both 1/f noise and white noise andas the voltage pump is a switched circuit it will also generate (highfrequency) switch noise. The switch noise is predominant at harmonics ofthe switch frequency of the oscillator. The switch frequency of theoscillator is designed to be larger than the upper cut-off frequency(e.g. 20 KHz) of the pass-band in the transfer function from themicrophone element to the microphone output terminal (T1). Typicaloscillator frequencies are 200 KHz, 500 KHz or 1 MHz.

The effect of all the noise sources of the charge pump can be diminishedby a low pass filter. This is the most efficient way of reducing theeffect of the noise. Reducing the 1/f and the white noise sourcesthemselves will require excessive area and/or current consumption. Thelow pass filter is embodied by means of the resistor 105, R and thecapacitor component of the microphone element 109. The cut-off frequencyof the low pass filter 105, 109 affects the pass-band in the transferfunction from the microphone element to the microphone output terminal(T1). Preferably, the cut-off frequency of the low pass filter 105, 109is substantially equal to a lower cut-off frequency (e.g. 20 Hz) of thepass-band in the transfer function from the microphone element to themicrophone output terminal (T1). The cut-off frequency of the low-passfilter 105 is preferably located at a frequency below the well-knownA-weighing curve.

In an embodiment the microphone element Cmic is coupled as anelectrically floating device. This is embodied by interchanging thecoupling capacitor 107 and the microphone element 109. It is within theskills of a skilled person to arrange the necessary change of thecircuit and terminals layout.

FIG. 2 shows a microphone comprising an integrated circuit with avoltage pump operated by a constant current drawing oscillator. In thisembodiment the oscillator 206 is configured to draw equal levels ofcurrent across multiple half-cycles of the oscillator signal provided bythe oscillator 206. Thereby, a more even and less noisy operatingcurrent is drawn through the combined microphone output and powerterminal Tc1 (T1).

The oscillator 206 comprises a constant current source 203 and a firstcircuit path 205 and a second circuit path 204 which alternately chargesand discharges a capacitor by alternately drawing current from theconstant current source 203. A voltage level sensitive element e.g. acomparator or Schmitt-trigger provides digital output oscillator signalsP1 and P2 to the voltage pump 104. A switch 207 is preferably applied todirect current from the constant current source 203 either through thepath 205 or through the path 204. The switch is controlled by respectivesignals from the paths 204 and 205. Preferably, the output signals P1and P2 are 180 degrees phase-shifted relative to each other.

Due to the constant current source 203 and the signal paths 204 and 205,the oscillator 206 is configured to draw substantially equal levels ofcurrent across consecutive oscillator signal half-cycles.

In an alternative embodiment only one of the paths 204, 205 isconfigured with a chargeable element e.g. a capacitor whereas the otherpath can be configured with a resistor. Thereby, a capacitor is savedwhen only a single pulse signal is to be provided to the voltage pump.

FIG. 3 shows a first oscillator with constant current consumption. Inthis embodiment, the oscillator comprises two current sources 303 and304 which are configured to draw a constant current from the powersupply, which can be provided via terminal Tps.

In this embodiment, the current sources 303, 304 are configured to eachdraw a constant current. The current sources are each coupled to power arespective inverter 301, 302.

Internally, the inverters are arranged to either draw the currentthrough an internal element, eg a resistor or transistor, or through itsoutput. Whether the inverter is in a state where current is drawnthrough the internal element or in a state where current is drawnthrough the output is controlled in dependence of whether the voltagelevel at its input is above or below a threshold voltage level. Thisthreshold can be set as a fraction of the reference voltage Vref. Theoutput of the inverters 301 and 302 is coupled to respective capacitorsC2 and C3. When the inverter 301, 302 is in a state where current isdrawn through the output, the capacitor is charged and the voltageacross the capacitor C2 and C3, respectively, will increase.Alternatively, in the other state of the inverter, the capacitor will bedischarged through the inverter or through another load.

A control circuit 305 is configured to provide input signals to theinverters 301 and 302. These input signals are provided as digitalsignals in response to the voltage levels at the output of theinverters, across the capacitors or as provided at terminals P1 and P2.The control signals are provided such that the capacitors C2 and C3 arealternately charged and discharged to provide oscillating signals P1 andP2 which are phase-shifted approximately 180 degrees relative to eachother.

Thus, at terminals P1 and P2 oscillating signals are provided which arefunctions of the charge on the capacitor C3 and C2, respectively. Toobtain digital signals threshold elements e.g. Schmitt-triggers can beprovided.

FIG. 4 shows a second oscillator with constant current consumption. Thisembodiment is an oscillator shown in greater detail. As it appears fromFIG. 3 the oscillator is built around two inverters 403 and 404. Theinverters 403 and 404 are powered by a current source T1 that is biasedby a bias circuit 402, Bias2 to make T1 provide a constant current.

The inverters are configured to either draw current through an internalelement, e.g. a resistor or transistor, or through its output. Whetherthe inverter is in a state where current is drawn through the internalelement or in a state where current is drawn through the output iscontrolled in dependence of whether the voltage level at its input,provided at circuit points ID1 and ID2, is above or below a thresholdvoltage level.

The output of the inverters 403 and 404 is coupled to respectivecapacitors C1 and C2. When the inverter 403, 404 is in a state wherecurrent is drawn through the output, the respective capacitor is chargedand the voltage across the capacitor will increase. Alternatively, inthe other state of the inverter, the capacitor will be dischargedthrough the inverter or through another load.

The voltage across the capacitors C1 and C2, dependent on their chargelevel, controls respective transistors T3 and T5. This is achieved by acircuit node connecting the gate terminals of transistor T3, capacitorC1 and output terminal of the inverter 403. Correspondingly, a circuitnode connecting the gate terminals of transistor T5, capacitor C2 andoutput terminal of the inverter 404.

Transistors T2 and T3 are coupled as constant current sources in serieswith transistors T3 and T5, respectively. The transistors T2 and T4 arebiased by a bias circuit 401, Bias1. T3 and T5 are controlled by thevoltage level across the capacitors C1 and C2 which in turn are chargedor discharged as determined by voltage levels at their input ID1 andID2. Thereby buffered oscillator signals P1 and P2 are provided.

The control circuit 405 is provided to control the circuit to provideout-of-phase oscillator signals P1 and P2. Preferably, 180 phase shiftedsignals are provided.

FIG. 5 shows a first embodiment of a first stage voltage pump. The firststage voltage pump comprises a buffer amplifier 501 which provides avoltage reference supply to voltage pumps 502 and 503. This referencesupply is regulated to be maintained at a fixed level determined by thereference voltage Vref input to the non-inverting input of the bufferamplifier 501. The output from the buffer amplifier 501 is provided as afeedback to the inverting input of buffer amplifier 501.

The voltage pumps 502 and 503 are preferably implemented by a stackedcapacitor principle. The voltage pumps 502 and 503 are connected toreceive oscillator signals from e.g. the oscillators shown in FIGS. 3and 4 in which the respective oscillator signals are designated P1 andP2. At their output, the voltage pumps provide signals P1′ and P2′. Thevoltage pumps can provide an increased voltage level of a factor 2 to 3e.g. 2.4 or higher, e.g. a factor of 4 to 10.

Normally any IC technology has a nominal operating voltage where allcomponents are specified to be operational. For most CMOS technologies3V or 5V are common. Below these nominal voltages all devices can beoperated safely without breakdown. Complex circuitry can be implementedhaving high performance.

Above this nominal voltage level a limited number of components arenormally available. That is, standard CMOS transistors can not be usedas they would brake down. However, there exist IC technologies withso-called High Voltage CMOS transistors, but these technologies areexpensive and High Voltage CMOS transistors are very bulky. Therefore,it is advantageous to divide the voltage pump into two sections: a lowvoltage part and a high voltage section. The low voltage sectionoperating below the normal operating voltage of the technology, normally3V or 5V and the high voltage section operating above. At the highvoltage section the IC components are configured precautionary to avoidbreakdown of the components at the High Voltage levels.

The signals P1′ and P2′ are thus pumped oscillator signals. Thesesignals are provided with regulated amplitude levels by means of thefeedback configuration. Thereby precise oscillator signals are provided.Since the feedback configuration requires a more complex circuit layout,all other things being equal, this first stage voltage pump ispreferably implemented on a portion of the chip die which is configuredfor operation at or below a nominal voltage level for that portion ofthe chip die and the amplitude levels of P1′ and P2′ is preferablysubstantially equal to the nominal voltage level for the portion. Afollowing voltage pump operating at the oscillator signals P1′ and P2′can be made simpler since precise amplitude levels are input to thefollowing pump. Additionally, since the amplitude of oscillator signalsare typically well below the nominal voltage level pumping of theamplitude to the nominal value reduces the number of following voltagepump stages to reach a given (pumped) bias voltage level.

FIG. 6 shows a second embodiment of a first stage voltage pump. In thisembodiment the output signals P1′ and P2′ are provided as feedbacksignals to the buffer amplifier 501 at its inverting input. The outputsignals P1′ and P2′ are provided as feedback via circuit 604 to obtainan accurate and/or fixed pulse level of the signals P1′ and P2′. Thecircuit 604 provides an output signal representing the amplitude of P1′and P2′ in combination e.g. by integration over P1′ and P2′.

In a preferred embodiment an additional feedback loop can be provided toobtain a desired stability of the circuit/regulator providing theaccurate and/or fixed pulse level.

FIG. 7 shows a third embodiment of a first stage voltage pump. In thisembodiment the inverters 502 and 503 which pump an input voltage e.g.2.4 or 3 times are replaced by an ordinary inverter which does not pumpinput voltage. However, in order to obtain the precise and relativelyhigh pulse amplitude level at the low voltage section, a voltage pump703 is provided. This voltage pump 703 provides a pumped supply voltageto the inverters 701 and 701. The inverters are configured to providethe oscillator signals P1′ and P2′ with pulse amplitude voltagessubstantially equal to the supply voltage of the inverters. Thereby analternative circuit for providing regulated oscillator signals isprovided.

In still an alternative circuit for providing regulated oscillatorsignals the feedback signal to the inverting input (−) of the bufferamplifier 501 is provided directly from the output of the voltage pumpVP, 703. Thereby, the output voltage provided by the inverters 701 and702 are not included in the feedback loop. But since the inverters arerelatively precise compared to the voltage pump 703 good regulation ofP1′ and P2′ is achieved. Thereby the circuit 604 can be avoided.

FIG. 8 shows a compound voltage pump. The compound voltage pump 707comprises a first stage voltage pump 802, UPC1 and a second stagevoltage pump. The second stage voltage pump comprises a cascade ofvoltage pumps 803, 804, 805, 806, UPC2.

The first stage voltage pump can be implemented in various ways, butpreferred embodiments of the first stage voltage pump have beendisclosed in the above. The first stage voltage pump is based on anoscillator 801 which provides oscillator signals P1 and P2 phase shiftedabout 180 degrees relative to each other. The oscillator signals areprovided to the voltage pump 802, UPC1 to provide pumped oscillatorsignals P1′ and P2′. It is recalled from the above that the pumpedoscillator signals are regulated to provide precise and at the same timerelatively high voltage levels. It is further recalled that the pumpedoscillator signals are provided by the circuits implemented in a lowvoltage section. This low voltage section is illustrated by the dashedbox 810.

If the pulse amplitudes of the repeated pulses constituting theoscillator signals P1′ and P2′ are maximized with respect to the nominalvoltage level specified for the low voltage section 810, the number ofcascaded voltage pumps at the second stage can be minimized, otherthings being equal. Consequently, a more chip area efficient design isprovided.

It is recalled that any IC technology has a nominal voltage at or belowwhich all components are specified to be operational without DC voltagebreakdown. At or below the nominal voltage complex circuitry can beimplemented with high performance. Above this nominal voltage level onlya limited number of components are available. That is, e.g. standardCMOS transistors cannot be used as they would brake down due to the highvoltage levels. The limited number of components comprise High VoltageCMOS transistors, but the technology for implementing the High VoltageCMOS transistors is expensive and the components are very bulky.Therefore it is advantageous to divide the charge pump into a lowvoltage section and a high voltage section.

Reverting to the description of the voltage pump: The pumped oscillatorsignals P1′ and P2′ are provided to each of the voltage pumps 803, 804,805, 806, UPC2 arranged in cascade. Each of the voltage pumps designatedUPC2 is provided with an input signal which at circuit nodes (b), (c)and (d) is characterized as a DC voltage superposed by an oscillatingsignal with a pulse amplitude largely about the pulse amplitude of P1′or P2′. The node (a) is preferably coupled to receive a DC signal fromUPC1. This DC signal can be a ground reference, a DC level e.g. the DCsupply voltage provided to the inverters 502, 503 or another DC signal.

The cascade of voltage pumps generates gradually larger voltage levelsfrom circuit node (a) to circuit node (b), to circuit node (c), to (d)and to (e). Each of the voltage pumps can add a voltage corresponding toe.g. four times the pulse amplitude of the oscillator signals to the DCsignal input to the voltage pump. However, this depends on theconfiguration of the pump and especially on the number of capacitors inthe configuration and the magnitude of loss in the pump.

The voltage level provided by voltage pump 805 at circuit node (e) isprovided via a series resistor 808, R and terminal Tc2 as a microphonebias voltage to provide an electrical charge on one of the microphonemembers.

The capacitor 809, C is coupled to block the pumped DC bias voltage fromreaching the input stage of a preamplifier (not shown) coupled toterminal Tc4 to receive a microphone signal from the microphone membercoupled to terminal Tc2 at which the bias voltage is provided.

The oscillator 801 and the voltage pump 802 is provided with operatingpower by drawing a current via terminal Tc5. However, the operatingpower could be provided via terminal Tc4 also providing the microphonesignal.

Especially for telecom microphones it is expedient to apply thismultistage voltage pump to obtain a relative large overall voltage pumpfactor per chip die area unit.

Preferably, the voltage pumps 803, 804, 805 and 806, UPC2 are of thesame type; preferably they are similar or identical.

High voltage IC components require larger mutual spacing, deeper wells,thicker gate oxide etc. That is, physically they are differentcomponents. In the following a voltage pump of the Dickson type forimplementation in the high voltage section is described.

FIG. 9 shows a second stage voltage pump. This voltage pump is shown inthe form of a Dickson-converter and constitutes preferably the modules803-806, UPC2 of the compound voltage pump. In this embodiment theDickson-converter comprises four diode-capacitor stages, but fewer ormore stages can be applied. The Dickson voltage pump usually consists ofseveral diode-capacitor stages. The numbers of sections depend on pulseamplitude of oscillator signals P1′ and P2′ and the desired outputvoltage. The voltage pump 901 receives an input voltage signal. In casethe pump 901 is coupled in a cascade the input signal can be provided bya preceding pump module as a DC signal superposed by an oscillatingsignal largely corresponding to P1′ or P2′. The input signal is providedat the terminal designated ‘In’ and provides a pumped output signal atits terminal designated ‘Out’. The pump is operated by the oscillatorsignals P1′ and P2′ to alternately charge the capacitors C1, C3 and C2,C4, respectively. When the voltage pump has reached a normal operatingstate and the pumped output voltage thus has reached a nominal level,each diode-capacitor stage adds a voltage step equal to the oscillatorpulse amplitude minus any loss at the stage. Consequently, an outputvoltage greater than the input voltage and the pulse amplitudes can beprovided.

The Dickson charge pump is characterized by a simple circuitconfiguration which can be implemented in many different ICtechnologies. The Dickson pump can be implemented using a very smallchip area. Additionally, the Dickson charge pump is characterized by lownoise and low output ripple when driving high impedent loads (amicrophone). That is, when the voltage pump has reached a normaloperating state and the pumped output voltage thus has reached a nominallevel, the current through the diodes D1, D2, D3 and D4 (or activedevices in the form of CMOS transistors etc.) becomes very small. Thisin turn increases the impedance of the diodes dramatically and thuseffectively filters any switching noise and noise from other circuitparts.

However, in the attractive and simple IC implementation its pump levelperformance is degraded by parasitic capacitors and its pump levelfactor relating the input voltage level to the output voltage level ispoorly controllable. Thereby, predictable or desired bias levels aredifficult to achieve. Since the multiplication factor depends onparasitic capacitors and because there is an absolute voltage dropacross each Dickson section as few Dickson stages as possible should beused to obtain an efficient circuit configuration.

The precision of the Dickson multiplier can be increased and the voltagedrop across each Dickson section can be minimized. But this is at theexpense of increased complexity and size. That is, the Dicksonmultiplier can then not be implemented using simple components but mustme implemented using high voltage transistors and a special high voltageprocesses. This causes larger area and higher cost.

Standard IC technologies, which are preferred, provide possibilities ofmaking components which can withstand higher voltages than the nominalvoltages. These components are based on parasitic capacitors and arethus not very precise and are poorly controlled. Note that metalcapacitors are based on parasitic coupling between metal layers,junction diodes in n-wells etc. But the use of such components ispossible only if a simple voltage pump, like the Dickson pump, isutilized. Use of the simple Dickson multiplier also enables us to use asubset of High Voltage components from High Voltage technologies e.g.deep N-Wells.

When the pulse amplitude of the oscillator signals P1′ and P2′ issubstantially equal to the nominal voltage of the low voltage section,the high voltage part can be configured with simple voltage pumps. Thisprovides the advantage of implementing the pump using parasiticcomponents which can withstand the high voltage. Examples of parasiticcomponents are metal capacitors, junction diodes and poly diodes.

FIG. 10 shows a second stage voltage pump with parasitic capacitors.This stage corresponds to the pump stage shown in FIG. 8, but hereparasitic capacities Cp1, Cp2, Cp3 and Cp4 are shown. The parasiticcapacities come into existence when capacitors C1, C2, C3 and C4 areimplemented on an integrated circuit chip.

FIG. 11 shows an IC implementation of a metal oxide capacitor.Generally, and when implementing capacitors of the voltage pump, MetalOxide Substrate (MOS) IC technology provides only limited capacity perarea unit. However, MOS technology is capable of withstanding highvoltages without voltage breakthrough (disruptive discharge). Thereforethe MOS technology is suitable for implementing high-voltage voltagepumps or cascaded stages of Dickson converters. It should be noted thatthe second stage of the compound voltage pump typically gives rise tovoltages above 10 volts, e.g. 15, 20, 30 or even up to 50 or 60 volts.These voltages are high voltages for technologies commonly available forconsumer product microphones such as mobile phones, cameras, PDA's etc.

The IC implementation of the metal oxide capacitor is sketched for acapacitor e.g. C1 of FIG. 10. The capacitor is implemented by means oftwo metal plates. The capacitor is implemented on a silicium substrate1101. On the silicium substrate a first layer of silicium oxide 1102 isprovided to isolate a first plate M1, 1106 electrically from thesubstrate. A second layer of silicium oxide 1103 is provided on thefirst layer and the plate M1, 1106 to isolate the first plate from asecond plate M2, 1105. A third layer of silicium oxide 1104 is providedto isolate the plate M2 from other circuitry or surroundings. Paths toconnect the capacitor plates are routed e.g. in the second and thirdoxide layers 1103 and 1104.

Since typically, the substrate is connected to a ground reference itappears that the substrate forms a capacitor plate which acts with thefirst plate of C1 to create the parasitic capacitor Cp1.

Compared to so-called poly-capacitors, the oxide layer is thicker in MOStechnology therefore the capacity is lower.

FIG. 12 shows an IC implementation of a poly-diode. IC implementation ofthe poly-diode is sketched for a single diode. The diode is implementedby means of layer sections of a P+ doted material 1203 and a N+ dotedmaterial 1204 arranged adjacent to the P+ section. The P+ doted and N+doted layer sections are arranged on a layer of silicium oxide toisolate the sections from the silicium substrate 1201.

Poly-diodes are expedient for voltage pumps since they can withstandhigh voltages. Typically, the oxide layer has a thickness of about 1000nm and can withstand up to 200-300 volts.

FIG. 13 shows an IC implementation of a diffusion diode in an N-well.The diffusion diode is placed in a well in a silicium substrate layer1301. The well 1304 is obtained by providing a N+ doted a portion of thesubstrate 1301. In the well 1304 a P+ doted portion 1302 is providedspaced apart from an N+ doted portion 1303.

The P+ doted portion is provided with a circuit path to provide anelectrical circuit node (1) and the N+ doted portion is provided with acircuit path to provide a circuit node (2). Thereby a diode D1 isformed. However, due to the physical structure of the N-well diode andsince the substrate 1301 typically is connected to a ground reference(node (3)), a parasitic diode D2 is also created.

FIG. 14 shows a charge pump with N-MOS switches. The charge pump is ofthe Dickson type and comprises four switch-capacitor stages: S1, C1; S2,C2; S3, C3; and S4, C4. The N-MOS switches 1401, 1402, 1403 comprises afirst N-MOS transistor 1401 coupled as a diode to provide bias of asecond transistor 1405 coupled as a controllable switch.

The switch S4 of the last switch-capacitor stage is coupled as diode.The capacitor 1413, CL represents a load (the microphone element). Thetransistor 1407 provides in combination with the capacitor 1412, C5control of the switch S3, 1403.

It should be noted that other types and implementations of a voltagepump are available.

FIG. 15 shows a compound charge pump with a master-slave configuration.In this configuration of the compound charge pump 807 a simple feedbackcircuit is provided in the high voltage section 811. The feedbackcircuit is based on a voltage pump module 1501 which is similar,preferably as similar as possible, to the voltage pump module 803. Thevoltage pump module 1501 receives the same inputs as the pump module803. The output of the pump module 1501 is coupled to a regulator 1502(shown in the form of a comparator). The output from the pump module1501 is compared to a reference voltage Vref and the regulator providesan error signal back to the voltage pump 802. The voltage pump 802 isconfigured to provide regulated amplitudes of the oscillator signals P1′and P2′ in dependence of the error signal. Thereby a master slaveregulating configuration is provided, where a voltage is sensed at thehigh voltage section, but at a relatively early stage of the cascadedvoltage pumps at the high voltage section.

In an alternative configuration the sensed voltage at circuit node (b)is provided directly from circuit node (a). Thereby, the master voltagepump 1501 can be saved.

FIG. 16 shows a microphone with an integrated circuit with a voltagepump and a preamplifier with a feedback filter. The voltage pump Vpmp,1601 provides a pumped voltage as a bias voltage Vb via the resistor Rc,1602. A capacitor Cc operates with the resistor Rc to decouple noisefrom the voltage pump. The voltage pump 1601 can be embodied as shown inFIG. 8, in which case the resistor Rc, 1602 corresponds to resistor R,808 and the capacitor Cc corresponds to the capacitor C, 809. However,in the shown configuration the microphone element Cmic, 1603 is coupledas a floating device. This configuration, of coupling the microphoneelement as a floating device, is especially expedient when themicrophone element is a MEMS device since MEMS microphones typicallyallow for being coupled as a floating device. The voltage pump receivesan operating current via the output terminal, designated Pwr/Out, of themicrophone 101.

In an alternative configuration the microphone element 1603 is coupledwith one of its microphone members to the ground reference cf. e.g.FIG. 1. For a typical electrostatic condenser microphone one of itsmicrophone members (typically the back plate) is configured to beconnected to the ground reference.

A non-inverting input (+) of the operational amplifier 1604 is coupledto receive a microphone signal from a first of the plate members of thecapacitor microphone element Cmic. The amplifier 1604 is provided with afeedback circuit 1605. The feedback circuit 1605 has an input portdesignated ‘a’ coupled to receive an output signal from the amplifier1604 and an output port designated ‘b’ coupled to an inverting input (−)of the amplifier 1604. The preamplifier comprising the amplifier 1604and feedback circuit 1605 is implemented on a semiconductor substrate1606.

The amplifier 1604 and the feedback circuit 1605 have, in combination, afrequency transfer function from the non-inverting input (+) to theoutput (which corresponds to the circuit node connected to the inputport ‘a’ of the feedback circuit). This frequency transfer function hasa high-pass characteristic. However, the feedback circuit has afrequency transfer function from the port ‘a’ to the port ‘b’ with azero and a pole; wherein the zero is located at a higher frequency thanthe pole. Thus the feedback circuit has a low-pass characteristic.

The feedback circuit in the form of a filter can be a first order filteror it can be of higher order; e.g. second order, third order or fourthorder. Also it can be implemented as a passive circuit or as an activecircuit. The feedback loop assures that the overall gain of theamplifier with feedback is relatively low at low frequencies (below e.g.20, 10 or 5 Hz) and relatively high at audio band frequencies (in therange e.g. 20 Hz to 20 KHz).

The preamplifier is powered from its output terminal designated Pwr/Out.However, it should be possible to draw a current through the terminalPwr/Out e.g. via a resistor 1608 coupled to a supply voltage Vdd. Theamplifier is coupled as a non-inverting amplifier with the microphoneconnected to the non-inverting input. This ensures that the capacitiveloading of the microphone is very low. Due to the feedback, theinverting input terminal (−) of the amplifier 1604 will exactly followthe non-inverting terminal (+). If the input stage of the amplifier 1604is a differential transistor pair (i.e. a differential stage),gate-source voltages of the transistor pair will remain constant and theinput capacitance will consequently be very low.

Reverting to bias voltage generated by the voltage pump: It is recalledthat when there is no electret layer on one of the microphone platemembers, an external bias is needed and can be supplied by a voltagepump integrated on the same semiconductor substrate as the preamplifier.Further, voltage pumps are normally quite noisy and thus a decouplingfilter is needed. This filter can consist of a decoupling capacitor Ccand a large resistor, Rc. To decouple the noise of the voltage pump1101, a filter with a very low cut-off frequency is needed. And thus itsettles very slowly during power up. That is, a very large low frequentsignal will be present on the input of the amplifier for a substantialperiod of time. So, the preamplifier with a low gain at low frequenciesproves to be very beneficial.

If the microphone is biased at a high DC voltage, a DC couplingcapacitance is needed between the amplifier and the microphone as theamplifier is in nearly all cases unable to handle the large DC levelwithout overload. Furthermore by integrating everything on the same chipthe total performance can be optimized yielding the best possibleperformance.

It should be noted that operating power to the integrated circuit canalternatively be provided via a separate power terminal, therebyproviding the microphone signal and receiving operating power viadifferent terminals.

FIG. 17 a shows a first feedback filter. The feedback filter 1605 formsa feedback circuit with an input port designated ‘a’ and an output portdesignated ‘b’. The input port, a, is connected to a ground referencevia a series connection of a first resistor R2, 1702; a capacitor C,1703; and a second resistor R3, 1701. The output port, b, is coupled tothe circuit node formed by the interconnection of the first resistor R,1702 and the capacitor C, 1703.

The feedback filter can be implemented in many ways, but not all of themare equally easy to integrate on a chip. Especially filter types withseries resistors are difficult to implement as the component valuesneeded are implemented with difficulty on a chip or semiconductorsubstrate.

The desired filter transfer function is a high-pass filter function.This is typically implemented using two resistors in series with acapacitor (see FIG. 17 a). At lower frequencies the transfer functionfrom port a to port b is close to one and at higher frequencies it isdetermined by the ratio of R2 and R3. In order to obtain low noise theresistors will have to be in the kOhm range and thus require thecapacitor value to be nF range to realize a desired cut-off frequency.Capacitors in the nF range would require excessive chip area, and suchsolutions are thus deemed to be not possible for a chip implementation.However, it should be within the skills of a person skilled in the artto modify the filter for expedient IC implementation.

FIG. 17 b shows a second feed-back filter for IC implementation. Thefeedback filter 1605 forms a feedback circuit with an input portdesignated ‘a’ and an output port designated ‘b’. The filter has aconfiguration with an input port, a, connected to a series connection ofa first resistor R2, 1707 and a second resistor R3, 1708 which forms aresistor node at their interconnection. The input port is also connectedto a series connection of a first capacitor C1, 1706 and secondcapacitor C2, 1704 which forms a capacitor node at theirinterconnection. The capacitor node forms the output port. Additionally,the resistor node and capacitor node are interconnected by a resistorR1, 1705.

For this configuration of the feed-back filter, the low frequencytransfer function from port a to port b is determined by the tworesistors R2, 1707 and R3, 1708. The high frequency transfer function isdetermined by C1, 1706 and C2, 1704. The cut-off frequency of the filtercan be set by R1, 1705. If R1, 1705 is chosen to be very large, thenoise of the filter will moved to very low frequencies and the audioband noise can thus be minimized without using excessively largecapacitor values. Suitable ranges for implementation on a semiconductorsubstrate are C1=1-500 pF, C2=1-500 pF and R1=GOhm-Tohm.

FIG. 18 shows a bootstrap configuration in connection with a detailedview of the preamplifier. The amplifier input stage 1801 comprises adifferential pair of PMOS devices 1803, 1806. This differential pairwill have to be optimized in both width and length as an optimum for 1/fnoise and white noise exists. If needed, an offset can be built into thedifferential pair by adjustment of the aspect ratio of the twotransistors in the differential pair. Alternatively or additionally, themirroring factor of the current mirror 1804, 1805 in the bottom of thedifferential stage can be adjusted. If the ratio between the aspectratio of the differential pair transistors are A and the current mirrorfactor is B, the offset of the amplifier will be n*Vt*ln(A*B). Where nis the so-called weak-inversion slope factor and Vt is a constantapproximately equal to 26 mV.

Various implementations of a differential input stage exist—forinstance, the NMOS current mirror 1804, 1805 can be replaced by aso-called folded cascode in combination with a PMOS current mirror.

At the output stage 1802 of the amplifier, an output transistor 1808 isconnected to the high impedent gain node. The function of this is to addgain and to isolate the high impedant node from the outside. Note thatthe only device which has a varying current is the output transistor.Thereby the other transistors are biased by constant current sources.

Thus, a principle amplifier with a differential input stage and anoutput stage is described.

As described above Metal Oxide capacitors inherently provide a parasiticcapacitor. Thus a DC blocking capacitor like capacitor C1, 107establishes a parasitic capacitor. This parasitic capacitor will degradethe microphone signal input to the preamplifier. However, as shown bycapacitor Cp, 1809 the effect of this parasitic capacitor can besubstantially removed by coupling the parasitic capacitor between theinput terminal of the amplifier at which the microphone signal isreceived (e.g. the non-inverting input) and the circuit node of theamplifier established between transistors 1807 and 1803 (1806). Betweenthese two nodes the transistor 1803 (1806 as the case may be) provides again of about 1 time (0 dB). Thereby the voltage swing across thecapacitor is kept at about 0 volt.

Alternatively, the parasitic capacitor is coupled between the invertinginput (−) and the non-inverting input (+) of the preamplifier. Therebythe voltage following property of the input terminals is utilized toreduce the effect of the parasitic capacitor Cp.

A first one of the plate members of the parasitic capacitor Cp is sharedwith the non-parasitic capacitor (e.g. the a DC blocking capacitor likecapacitor C1, 107). The other plate member of the parasitic capacitor Cpis basically the substrate of the IC, but since this is coupled toground, a shield between the substrate and the first plate member isprovided to establish a contact to the parasitic capacitor. The shieldis provided by means of e.g. an N-well diode which provides forelectrical insulation to the substrate while being a plate member of theparasitic capacitor.

FIG. 19 shows a first embodiment of a microphone comprising anintegrated circuit with a sigma-delta A/D converter. The sigma-delta A/Dconverter 1903 receives a clock-signal via a terminal Tc4 of the chipdie 1902 and terminal T3 of the microphone 1901. In response to theclock-signal, serial digital data are outputted via terminals Tc5 andT4. The sigma-delta converter receives a voltage reference from theregulator 1904 which also provides a voltage reference or supply to theoscillator 103 and to the amplifier 108. The regulator 1904 receivesoperating power via separate terminal Tc6 (T5).

The low pass filter 904, LPF filters the pumped voltage from the voltagepump 104 to diminish primarily switching noise in the pumped voltagesignal.

FIG. 20 shows a second embodiment of a microphone comprising anintegrated circuit with a sigma-delta A/D converter. In this embodiment,the clock-signal that controls the sigma-delta converter 1903 is alsoapplied as an input oscillator signal to the voltage pump 103.Preferably, the voltage pump provides a 180-degrees phase shiftedversion of the clock signal to provide signals P1 and P2 for subsequentprocessing as described in the foregoing.

In an alternative embodiment a frequency divider or multiplier isprovided to shift the oscillator frequency relatively between thesigma-delta converter and the voltage pump. The shared use of theexternally provided clock-signal reduces the chip die area consumption.

Generally, it should be mentioned that the microphone alternatively canbe coupled in a floating configuration. For instance, the microphone,Cmic, can replace the capacitor C1 or 906. The capacitor C1 or 906 canaccordingly replace the microphone, Cmic.

Generally, in connection with preamplifier configurations it is notedthat the preamplifier can be of different, types. The preamplifier canthus be implemented as a differential amplifier providing a differentialoutput signal. Thereby, a signal provided in response to a soundpressure on the microphone is provided to a the differentialpreamplifier. The differential preamplifier is characterised by having again characteristic with relative low gain for frequencies below anaudible range and a relative high gain for frequencies in the audiblerange. Preferably, the gain characteristic descent as a 1^(st), 2^(nd),3^(rd), 4^(th), or higher order below the audible range. In additionthereto the amplifier is characterised by processing a low frequencymicrophone signal as a common-mode signal and a high frequencymicrophone signal as a differential mode signal. Thereby low frequencycomponents are effectively suppressed. The differential output signal isprovided across two terminals as a microphone preamplifier outputsignal.

Especially in connection with sigma-delta modulators it is important toremove low frequent signals since otherwise the low frequent signalswould generate idle-mode tones in the digital signal provided by thesigma-delta converter.

FIG. 21 shows a microphone cartridge. The microphone cartridge 2101 isshown in a schematic view illustrating an integrated circuit IC, 2102and a microphone element 2103. The integrated circuit 2102 comprises thepreamplifier and voltage pump as disclosed above and is embodied on asemiconductor substrate or a single chip die.

FIG. 22 shows a schematic view of a microphone with an integratedcircuit and a MEMS microphone element. The microphone 2201 comprises aMEMS microphone member integrated on a first substrate 2202. On a secondsubstrate 2203 the preamplifier circuitry and the voltage pump isprovided as described in the foregoing. The first and second substratesare bonded to each other to provide electrical connections.

The preamplifier circuitry comprises one of the different embodimentsdisclosed above i.e. comprising a voltage pump and a preamplifier.Optionally, the preamplifier is provided with a feedback circuit toprovide a preamplifier with a high pass filter characteristic.

It should be noted that the MEMS microphone element, the voltage pumpand the microphone preamplifier can be integrated on a singlesemiconductor substrate.

In a preferred embodiment the input side of the preamplifier is coupledto a limiter circuit. When the charge-pump starts up a very large signalwill be present at one side of the microphone capacitor. Without alimiter, this large signal will also couple to the input of thepreamplifier. This signal has a large low frequency content and willthus overload the preamplifier if it has a nominal gain at lowfrequencies. Such a “power on” pulse from the charge pump will beequivalent of a very large input sound pressure level. A limiter at theinput of the preamplifier can be designed to clamp the start up pulsewhile largely not affecting normal operation. Such a limiter can beimplemented by means of two diode coupled transistors or a pair of backto back coupled diodes. If the preamplifier additionally is designed tohave a gain vs. frequency function which has a high pass filter functionthen the identified problem can easily be avoided.

Integrated circuits are characterized from discrete circuits byphysically being implemented in the same silicon.

Generally, a microphone based on the principle of a capacitor, where oneof the capacitor plates is a moveable membrane without an electret layer(or substantially without an electret layer) is denoted a condensermicrophone.

The above features may be applied in embodiments of a preamplifierconfiguration that comprises a gain stage with a feedback filter, wherethe configuration has a relatively low gain response for frequenciesbelow an audio band and has a relatively high and substantially flatgain response in the audio band. The audio band can be defined to be anyband within the typical definition of an audio band. A typicaldefinition can be 20 Hz to 20 KHz. Exemplary lower cut-off frequenciesfor an audio band can be: 20 Hz, 50 Hz, 80 Hz, 100 Hz, 150 Hz, 200 Hz,250 hz. Exemplary upper cutoff frequencies the an audio band could be 3KHz, 5 KHz, 8 KHz, 10 KHz, 18 KHz, 20 KHz. By substantial flat is meantgain response variations within approximately +/−1 dB; +/−3 dB; +/−4 dB;+/−6 dB. However, other additional values of variation can be used todefine the term ‘substantial flat’.

In the above different preamplifier configurations have been disclosed.These configurations comprise different input/output terminalconfigurations e.g. a two-terminal configuration. However, it should benoted that three, four or more terminals can be provided forinput/output of signals to microphone and preamplifier. Especially, itshould be noted that separate terminals can be provided for supplyvoltage (at a first terminal) and preamplifier output (at a secondterminal). In case of a differential preamplifier output two terminalsfor the output signals can be provided in addition to a terminal forpower supply.

A separate terminal is provided for a ground reference. This groundreference is typically, but not always, shared by the power supply andoutput signal.

1. An integrated circuit configured to provide a microphone outputsignal, comprising: a preamplifier coupled to receive an input signal,generated by a first microphone member that is movable relative to asecond microphone member; and a voltage pump to provide a bias voltageto either microphone member.
 2. An integrated circuit according to claim1, where the integrated circuit is configured with an oscillator drivenvoltage pump to provide a bias voltage to either microphone member; andwhere the oscillator is configured to draw substantially equal levels ofcurrent across signal cycles provided by the oscillator.
 3. Anintegrated circuit according to claim 1 or 2, where the oscillatorcomprises paths with elements that can be charged with an electricalcharge and where the paths are controlled by the oscillator to chargethe different elements of the different paths alternately by a currentdrawn from a common source.
 4. An integrated circuit according to any ofclaims 1 to 3, where the voltage pump has a first pump stage at which anoscillating signal with a voltage pulse level is pumped to a highervoltage pulse level, and a second pump stage at which a voltage level ispumped to a higher level by means of a circuit operating on theoscillator signal, provided at the first stage, with the higher voltagepulse level.
 5. An integrated circuit according to any of claims 1 to 4,where the integrated circuit comprises a first portion configured with acircuit component layout for electrical operation at or below a nominalvoltage level and a second portion configured with a circuit componentlayout for electrical operation above the nominal voltage.
 6. Anintegrated circuit according to any of claims 1 to 5, where an outputsignal of the first voltage pump stage is provided as a feedback signalto a circuit which maintains a fixed voltage pulse level of the signalsoutput from the first pump stage (P1′; P2′).
 7. An integrated circuitaccording to any of claims 1 to 6, where the voltage pump has a firstpump stage providing an intermediate bias voltage and a second pumpstage providing the bias voltage from the intermediate bias voltage; andwhere the second pump stage comprises a voltage pump configured as aDickson converter.
 8. An integrated circuit according to any of claims 1to 7, where an output signal of the voltage converter of the Dicksontype is provided as a feedback signal to a circuit which provides aregulated voltage pulse level of the signals output from the voltageconverter.
 9. An integrated circuit according to any of claims 1 to 8,where multiple voltage converters are cascaded to provide the biasvoltage, and where a further voltage converter, which matches the firstconverter in the cascade, is coupled to receive the same signal as thefirst converter and to provide a feedback signal to a circuit whichmaintains a fixed voltage level of the signals output from the furthervoltage converter.
 10. An integrated circuit according to any of claims1 to 9, where the voltage pump comprises capacitors implemented as Metalcapacitors.
 11. An integrated circuit according to any of claims 1 to10, where the voltage pump comprises diodes implemented as Poly-diodes.12. An integrated circuit according to any of claims 1 to 11, where thevoltage pump comprises diodes implemented as diffusion diodes in anN-well.
 13. An integrated circuit according to any of claims 1 to 12,where the preamplifier, comprises a differential input stage with afirst and a second input terminal and an output stage with an outputterminal; a feedback circuit, with a low-pass frequency transferfunction, coupled between the output terminal and the first inputterminal and integrated on the semiconductor substrate; and where thesecond input terminal provides an input for a microphone signal.
 14. Anintegrated circuit according to claim 13, where the feedback circuit isa filter with a transfer function, in the frequency domain, with a zeroand a pole; wherein the zero is located at a higher frequency than thepole.
 15. An integrated circuit according to claim 13 or 14 where thepreamplifier has a transfer function, in the frequency domain, with azero and a pole; wherein the pole is located in the range 0.1 Hz to 50Hz or 0.1 Hz to 100 Hz or 0.1 to 200 Hz.
 16. An integrated circuitaccording to any of claims 13 to 15, where the feedback circuit is afilter which, in the frequency domain, has a relatively high gain levelbelow a transition frequency range and a relatively low gain level abovethe transition frequency range.
 17. An integrated circuit according toany of claims 13 to 16, where the transition frequency range is locatedbelow a frequency of about 100 Hz.
 18. An integrated circuit accordingto any of claims 13 to 17, where the transition frequency range islocated below a frequency of 40 Hz.
 19. An integrated circuit accordingto any of claims 1 to 18, comprising a DC blocking capacitor coupled todiminish a DC voltage at the input of the preamplifier, which DC voltageoriginates from biasing the first or second microphone member.
 20. Anintegrated circuit according to any of claims 1 to 19, where theintegrated circuit comprises an A/D converter.
 21. An integrated circuitaccording to any of claims 1 to 20, where the integrated circuit furtheris configured with an analogue-to-digital converter; and wherein thevoltage pump and the analogue-to-digital converter are driven by acommon clock-signal.
 22. An integrated circuit according to claim 20 or21, where the analogue-to-digital converter is of the sigma deltaconverter type.
 23. An integrated circuit according to any of claims 1to 22, comprising a high-pass filter.
 24. An integrated circuitaccording to any of claims 1 to 23, where the preamplifier is configuredto provide a high-pass filter function.
 25. A microphone comprising anintegrated circuit according to any of the claims 1 to
 24. 26. Amicrophone according to claim 25, where the microphone is a condensermicrophone.
 27. A microphone according to claim 25, where the microphoneis a MEMS microphone.